Superimposed high striking voltage circuit



Jan. 11, 1966 5, 555 3,229,159

SUPERIMPOSED HIGH STRIKING VOLTAGE CIRCUIT Filed May 16, 1960 5 Sheets-Sheet 1 IN VEN TOR.

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United States Patent 3 SUPERIMPOSED HIGH STRIKING VOLTAGE CIRCUIT Robert S. Webb, Bloomfield Hills, MiclL, assignor to Elox Corporation of Michigan, Troy, Mich, a corporation of Michigan Filed May 16, 1960, Ser. No. 29,363 22 Claims. (Cl. 315171) a This invention relates toimprovements in methods and apparatus for electrical-discharge-machining, sometimes referred to as E.D.M., arc-machining, or spark-machining, and this application is a continuation-in-part of my copending application Serial No. 747,078, filed July 7, 1958, now abandoned.

During recent years, the electrical-discharge-machining process has been used increasingly in the forming of cavities in very hard materials such as tool steels, cemented carbides, and the like. Improvements have been made in rate of machining, accuracy and finish, and in practically all of the modern E.D.M. apparatus now in use, electron tubes are utilized to obtain the rapid .interruption of the power circuit that is required for rapid stock removal with good surface finish.

Electron tubes commercially obtainablevare severely limited in their power carrying capacity. These devices are high-voltage, low-current devices. The machining gap in E.D.M. apparatus, on the otherhand, has a voltage drop of only about 15 volts. The present method of achieving high machining rate is to pass as high as possi- 3,229,159 Patented Jan. 11, 1966 In the drawings:

.FIG. 1 is a schematic wiring diagram of a typical E.D.M. power supply constructed in accordance with my invention;

FIG. 2 is a graphical representation of the grid drive voltage of the power tube bank in the above power supply;

FIG. 3 is a similar representation of the voltage in the primary of the power transformer;

FIG. 4 represents the voltage in the secondary 18 of the power transformer;

FIGS. 5, 6 and 7 are similar representations of a similar set of conditions, but showing a longer on time pulse;

FIG. 8 shows a modification of the power supply circuit wherein a voltage doubler circuit connects the power source to the machining gap; I

FIG. 9 is a modified circuit designed to operate at high current but low frequency and useful principally for roughing purposes;

FIG. 10 is a modification of the circuit utilizing transistors instead of electron tubes;

FIG. 11 is a further modification showing the superimposed high striking voltage principle applied to a relaxation oscillator circuit;

FIG. 12 is a modification of the FIG. 1 circuit in which a current limiting network is used;

FIG. 13 is a graphical representation of the voltage wave form of the FIG. 12 circuit; and

ble current through the gap which necessitates paralleling tubes in banks, sometimes hundreds in number.

For example, in one machinecurrently in use, a bank of 150 type 6AS7 vacuum tubes connected in parallel comprise the power supply to the machining gap. A 115 volt input suply is connected to the machine and the circuit interruption characteristic. is such that power pulses are delivered to the gap approximately one-third of the time. The peak current'is about 150 amperes and the average current about 50 amperes, the voltage drop through the power circuit being about 100 volts. It is known, however, that 6AS7 tubes and some other types are capable of interrupting circuits with voltages much higher than 115 volts.

Accordingly, it is an object of my invention to provide an improved E.D.M. circuit wherein much higher currents are delivered to the machining gap with the same number of vacuum tubes and with substantially the same type of interruption circuit as is now in use.

Another object is to increase the overall power efficiency FIG. 14 is a similar graph of the FIG. 8 circuit.

It will be understood that in the apparatus about to be described, automatic servo power feed of the machining electrode is contemplated, in accordance with present day practice in the E.D.M. art. The details of such a power feed have been omitted in the interest of brevity, and reference is made to my copending application, Serial No. 805,989, filed April 13, 1959, and Serial No. 15,505, filed March 16, 1960, for examples of power feeds suitable for use with the apparatus herein described.

Referring to FIG. 1, it will be seen that I have shown at 10 the main power supply for the apparatus, which I may be used for more delicate machining, particularly by a very substantial amount and to make possible utilization of the full voltage carrying characteristics of the tubes.

A further object is to effect a decrease in the bulk and cost of E.D.M. power supplies for given requirements.

The principal object is to provide in apparatus of the aforesaid type, means for superimposing on the machining voltage pulse a voltage pulse of much higher magnitude thereby providing increased overcut where a particular increase in gap size is desired, and improved power feed stability due to the larger gap clearance.

A still further object is to provide a machining circuit utilizing transistors instead of electron tubes which incorporates the above advantages.

Another object is to provide an inexpensive thyratron circuit for superimposing the above mentioned voltage pulse of much higher magnitude.

Other objects and advantages will become apparent from the following specification which, taken in conjunction with the accompanying drawings, discloses preferred forms of my device.

finishing operations.

An additional secondary winding 19 of relatively low power, high voltage characteristic is connected in parallel with the machining gap 21 through a rectifier 23 and a resistor 25.

It is, of course, understood that the gap 21 is supplied with suitable coolant in accordance with standard E.D.M. practice.

The other side of primary 14 is connected to the anode 20 of a power tube 22. It will be understood that the tube 22 represents a bank of tubes (in this instance 6AS7s) connected in parallel. Almost any number of such tubes may be so connected to provide the required power flow through the gap.

The secondary 18 of the power transformer 16 is connected at one side to the electrode 24, and at the other side to a workpiece 28 through a blocking diode 26.

The power tube bank 22 is controlled by a multivibrator network which comprises tubes 36 and 38. These tubes are preferably pentodes, type 6DQ5. The plates or anodes of these tubes are connected through load resistors 40, 42, and lead 48 to the positive terminal of a suitable power supply 44, the negative terminal of which is connected with the cathodes of the tubes by lead 46. The power supply '44 may be separate or it may be derived from the main supply 10 as desired.

The control grids 50, 52, of the tubes 36, 38, are crossconnected to the anodes 54, 56, respectively through coupling condensers 58, 60, andare connected to the positive side of the multivibrator power supply through the grid resistors 62, 64.

The output signal from multi-vibrator tubes 36, 38, is fed into an amplifier, which may comprise one or more pentode tubes 66, through condenser 68 and clamped to negative bias voltage 70 through diode 72. The amplitied and resquared signal from tube 66 is fed to the grid 74 of pentode 76 (which may be one of a bank) Where it is again amplified before being fed to the power tube bank 22'. The coupling to the driver tube 76 is through a coupling condenser 78 and a clamping diode 80 is provided to insure positive cut-off characteristic. Suitable isolation and signal resistors are also provided as shown to control the operating characteristics of diodes 72 and 80.

The power required to drive the main power tube bank 22 is in the order of several hundred watts, and to obtain increased efliciency, the amplifier 76 is floated in the grid circuit of the bank 22 rather than connected to the negative terminal of bias supply 82 as would be expected. Since the control signal appears between the cathode of driver 76 and point 84 of the circuit which is grounded, the network just described, which comprises a multivibrator and two stages of amplification, may be thought of as a floating signal source. 1

The output signal from this network is of rectangular wave form and is of substantially greater magnitude than that obtained from the conventional square wave generator. Normally these signal generators have an output ofapproximately ten watts. In the E.D.M. circuit of FIG. 1,'the' power required to drive the grids of the tube bank 22 is in the order of two hundred watts and more. A booster power supply 86 is preferably provided in series with the bias supply 82 to provide adequate voltage for the plate 88- of driver 76.

The output signal from driver tube 76 is developed from the voltage drop across variable resistor 90, which signal pulse with the added voltage of power source 82 constitutes the drive to the grids 92 of the bank 22. Proper adjustment of the circuit parameters will provide a signal at grids 92 having a selected on-time characteristic such as indicated in FIGS. 2 and 5, which illustrate graphically two somewhat extreme conditions.

As stated above, the signal generator power supply is the source; 44. Resistors 94 and 96, the latter being shunted by a condenser 98, are provided as shown.

The primary 14 of a transformer 16 has a damping network consisting of diode 100, resistor 102 and shunt capacitance 104 connected in shunt therewith.

The transformer 16 must be a stepdown transformer capable of handling relatively high currents at relatively high frequencies. The development of extremely thin iron lamination stock and specialized design now makes possible the design of transformers having the characteristics required for the circuit of FIG. 1. The transformer selected should have a maximum voltage swing on the primary equal to the peak voltage rating of the power tube selected and a turns ratio which will match the gap voltage required in E.D.M.

The aforementioned damping network limits the induced voltage or negative fly-back in the primary 14, which occurs between power pulses, to the voltage rating of the tubes 22 and this prolongs the lives of these tubes.

As so far described, it will be seen that the, tube bank 22 normally is biased to non-conducting condition by voltage source 82. An amplified signal from the multi vibrator will be impressed on the grids 92 of the power bank 22 and will overcome the normal grid bias and render the tube bank conductive. -In accordance with the preselected adjustment of the circuit parameters, a voltage will occur across the primary 14 as graphically repre- 4 sented (for example) by FIG. 3, which will induce a voltage in the secondary 18 like that represented in FIG. 4. This secondary voltage is instantly effective across the gap between electrode 24 and workpiece 28, and a power pulse will be delivered across the gap eroding the workpiece. At the same instant, the full voltage of secondary 19 is applied to the gap in parallel with the pulse from winding 18. Once the gap is fired, the current buildup. will, for all practical purposes, cut winding 19 out of the circuit because of the loading of the resistor 25. Substantially all of the power to the gap will be delivered by winding 18. The characteristic of winding 19 may be 7 chosen to provide almost any desired striking voltage .as

this higher voltage is blocked from winding 18 by rectifier 26. This permits the latter Winding to be designed for optimum power delivery to the gap. This sequence is repeated at high frequency until the machining operation is" completed or the operation interrupted by the machines power feed, as is known in the art.

.The gap between electrode 24 and workpiece 28 is flooded with dielectric fluid during machining as is com:

mon in E.D.M. x The circuit of FIG. 1 includes a fwatch-dog which functions automatically to cut-oif the power to the gap mally is biased non-conducting by the shnntresistor and condenser network 114, 116, which is connected across the voltage source. 82 through the screen voltage resistor 118 and the voltage reducing resistor 120. The voltage across resistor 90 plus that of the source 82 is, of course, the voltage which drives the grids 92 of the power tube bank 22. A selected portion of this voltage is thus effective on the grid 108 of cut-off tube 106 and tends to render tube 106 conductive whenever bank 22 is rendered conductive. The plate of tube 106 is connected to the grid circuit of multivibrator tube 38 by line 107 and conduction through tube 106 will instantaneously cut-off operation of the multivibrator.

However, the secondary of a transformer 122 (called for convenience the cut-01f transformer) is connected across the resistor .110 through a blocking diode 124.

The primary of the transformer 122 is connected across the gap between electrode 24 and workpiece 28 through a limiting resistor 126.

If the apparatus is functioning normally, a drive sig-' nal on grids 92 of the bank 22 will result in a striking voltage appearing across secondary 18 of power transformer 16 and the gap will fire. "This voltage would have to be only about 20 if there were no losses in the firing circuit. However, normal circuit losses require a voltage magnitude of 60 volts or more, and should a short circuit occur across the gap, the short circuit current would be almost of normal. With narrow pulse operation, as graphically illustrated in FIG. 4, the peak current selected is usually the peakpulse rating of the individual tubes of the power tube bank, anda 150% overload of this pulse current would strip the tube cathodes with comparatively sfew pulses. Thus ordinary short circuit cut-off devices, such as thermally responsive devices, operate too slowly to provide protection.

My per-pulse cut-oil device permits the power circuit to be operated with maximum 'efliciency because it renders it unnecessary to limit the power input to the gap to less than maximum desired on account of possibility of short circuit. The cut-off device operates to cut off the power input instantaneously, :that is to say, in about 5% of the period of a power pulse, and thus provides complete safety to the apparatus. This cut-cit device is extremely important in the operation of the machine especially when precision machining of expensive workpieces is being performed, where heat checking of the hole being cut might require scrapping of the piece. The readiness of the device to function instantly is constantly maintained by the precise balancing of the circuit parameters. The connection of grid 108 to the keying resistor 90 tends to render tube 105 conductive each time the multivibrator pulses, but the dominating negative bias of the network 114-116 inhibits conduction of tube 106 in the absence of any keying signal. During normal operation, the keying pulse voltage across resistor 90 is exactly neutralized in the grid circuit of tube 196 by the action of circuit 122, 124, 110'. However, appearance of a voltage across primary of transformer 122 (gap voltage) lower than a preset minimum will upset this voltage balance and instantaneously cause tube 106 to conduct and cut off the multivibrator through line 107. It is, of course, clear that the leading edge of the power pulse just initiated will cross the gap, but the cut-off is so fast that the power pulse will be literally squelched after initiation and no appreciable power will be delivered to the gap.

Interruption of operation of the multivibrator will, of course, cut off tube bank 22 as well as tube 106. After the normal pulse repetition delay time, the multivibrator will resume pulsing, and if the trouble in the gap which caused the abnormal low voltage has cleared, such as by back-up power feed, clearing of sludge, or the like, normal machine operation will be restored automatically.

It will be understood that the cut-off circuit shown is notlimited to use with the particular power delivering circuit shown. It would be equally useful with other gap power circuits whether of the impedance matching type or not.

Reference is now made to FIGS. 2, 3 and 4, which show graphically voltage conditions in certain portions of the FIG. 1 circuit under one selected set of conditions. FIG. 2 shows the grid drive voltage on the grids of power tubes when a signal of relatively short on time" per cycle is received from the multivibrator. The point A of FIG. 2 represents the negative grid bias normally impressed on the grids 92. This negative voltage is elfective on the grids for portions of the cycle represented by the lines AB and EF. The curve BCDE shows that the grid voltage is rendered positive by at least a sufficient amount to render the tube bank conductive for a period CD, the grids being made negative again, as indicated by DEFG for the remainder of the cycle. FIG. 3 shows that in response to the short pulse received from the power tube bank, a voltage AB is impressed on the primary of transformer 16 for a time BC. FIG. 4 shows the voltage pulse ABCD delivered to the gap between electrode 24 and workpiece 28, the negative flyback of the secondary winding DEFG being blocked from the gap by rectifier 26. There cannot be, therefore, any reverse polarity rpulse across the gap.

FIGS. 5, 6 and 7 show a set of conditions similar, re-

spectively, to FIGS. 2, 3 :and 4, except that the primary voltage pulse triggered by the multivibrator is of rel tively long duration.

In any event, for successful normal operation, the secondary voltage of correct polarity to fire the gap must be of sufficient magnitude to deliver on open circuit enough power to achieve a striking voltage at the gap of at least thirty volts and a sustained voltage in the 'order of twenty volts, taking into consideration the with the bulk of the machining power being delivered at low voltage but at high current. Thus, extremely good economy is achieved, both as regards electronic circuitry and power input.

For a more detailed consideration of the power pulse delivered by the secondary 18, reference is again made to FIG. 3. It is assumed that the transformer 16 has a 5 to 1 ratio, approximately 300 volts being impressed on the primary from the tube bank 22 and 60 volts being available across the secondary 18. The current amplified pulse is indicated by the rectangular wave curve ABCD, which pulse is of correct polarity and power phasing to deliver power to the gap. Flyback voltage DEFG is effectively blocked by rectifier 26 to prevent gap discharge of opposite polarity.

Reference is made to FIG. 12 which shows a modification of the circuit in which a resistor is connected across the machining gap, a condenser-limiting resistor network 132-184, is connected in series with a diode 180, to the high striking voltage winding 19'. This circuit provides for increased current flow from winding 19 during the initial portion or leading edge of a particular power pulse. Thus a relatively low impedance striking pulse is delivered which quickly ionizes the gap and forces conduction, the additional power from condenser 184 assisting.

In this circuit, the resist-or 182 may be of relatively high value thus achieving increased efficiency during the machining portion of the cycle, since little power is drawn through the resistor 182 thereby allowing increased power to flow from the more efiicient low voltage winding 18'. In addition, the average or RMS wave form impressed across the gap is substantially less and consequently the hazard to the operator and to the equipment is less.

This may be better appreciated by analyzing the voltage wave forms shown in FIG. 13. Here, the open circuit voltage magnitude of winding 18', and of winding 18 (FIG. 1), is shown at 185. The open circuit voltage magnitude of Winding 19 (FIG. 1) is shown at 187, this voltage being effective across the gap during the entire positive or in phase portion of the cycle. However, the open circuit gap voltage produced by network 186, 182, 184 (FIG. 12) is shown at 189. A high-magnitude spike occurs at the leading edge of the pulse for the duration of time determined by the network 182-184, and quickly falls to the value 185 of winding 18'.

The latter wave form has a much lower RMS value and therefore eliminates to a large extent the hazard of the high voltage during this condition of open circuit.

A typical cutting pulse is indicated at 191, during which the very high striking voltage at the leading edge of the pulse quickly drops to the typical 15-20 volt level of the E.D.M. gap conduction voltage. The balance of the voltage of winding 18' is dissipated across resistance 193 and inductance 195 which represents the lumped losses inherent in the circuit.

The important factor in the operation of these circuits is the very high striking voltage which initiates each gap discharge and thus make possible use of a wider gap which improves power feed stability and makes possible large overcut when such is required.

Referring now to FIGS. 8 and 14, it will be seen that I have applied the principle of the voltage doubler circuit to produce a high-voltage, low power striking voltage magnitude.

If, in a normal machining pulse cycle, as illustrated graphically in FIG. 14, the portion 195 is long compared to the off-time portion 197, the magnitude of the negative portion of the pulse will be substantially greater than the positive portion. This flyback voltage is stored across the relatively small condenser 160 (FIG. 8) which is connected across the gap through diode 158. The condenser 16@ need be only of sufficient size to supply the power required to ionize the gap.

During the machining portion of the cycle, the component 1% is generated across the transformer and is in phase with conduction through the power tube bank. At this time, the voltage stored in condenser 160 is of proper polarity and is superimposed upon the machining pulse. Diode 158 blocks reverse discharge and this greatly increased voltage is effective across the gap. If the gap ionizes and conduction results, the condenser 160 is rapidly discharged through the transformer winding and gap and the bulk of the machining power flows through diode 162 in phase with pulse tube bank conduction.

This voltage doubler circuit is satisfactory as a superimposed striking voltage source where relatively small increase in magnitude of striking voltage is required, or where relatively long on time occurs with respect to off time during a machining pulse cycle, as shown in FIG. 14.

It must be understood that the circuit shown in FIG- URE 8 including the vacuum tube in the primary of the transformer is shown as a modification of the circuitry of FIGURE 1. Since one vacuum tube bank only is used, this circuit is known in the trade as single ended and as such is capable of delivering substantial power only in phase with vacuum tube conduction. If in this circuit condenser 160 becomes excessively large, the flyback voltage is excessively loaded causing an excessive component of DC. flux in the core, thus destroying the pulse characteristics of the transformer.

It is found, however, that the load presented by condenser 160 may approach of the total without destroying the pulse characteristics of the transformer.

This superimposed voltage of approximately 10% maximum power is, however, sutlicient to ionize the gap and permit operation in accordance with the principles set forth herein.

If a source of A.C. is used in place of the vacuum tube and transformer of if a double ended or push pull transformer circuit is used, condenser 160 may be large and thereby utilize both portions of the A.C. wave form.

Reference is now made to FIG. 9 which shows a form of circuit particularly suitable for roughing purposes. Here, single phase A.C. line voltage is used for input and the output to the gap is of low frequency, 60 cycles for example.

A stepdown transformer 204 has a primary winding 206 connected across line terminals 208, 210, and a secondary 212 across which is connected a full wave rectifier 214. The latter has balancing resistors 216 across each diode. The negative output of this rectifier is connected to the workpiece 200 in this instance, and the positive output is connected through resistor bank 272 to the machining electrode 202. The flow of gap current is controlled by the number of resistors 220 in the circuit, switches 222 being provided for controlling the bank. Note that resistor 218 is permanently connected, thus providing low loop impedance at all times. The output of this network is full wave rectified A.C. which presents to the gap 120 cycle pulsating DC. power.

A trigger circuit for providing high striking voltage is associated with this low frequency supply. Connected directly to the single phase input lines 208, 210, is a full wave rectifier network 224 having in its output a current limiting resistor 266 and storage condensers 226. Also connected across the input line is a transformer 228 having a primary 230, and secondaries 232, 234, 236.

Secondary 232 supplies filament voltage to trigger thyratron 256. Winding 234, in conjunction with diode 238 and current limiting resistor 240, provides DC. bias across condenser 242 and resistor 244 for maintaining thyratron 256 normally non-conductive.

Secondary 236 of transformer 228 is connected to full wave rectifier 246 which has in its output saturable transformer 252, series connected with a network consisting of resistor 248 in parallel with condenser 250. Transformer 252 is wound of oriented sharp knee iron such that it has a very sharp saturation characteristic and is provided with sufficient primary turns that it saturates readily during only a portion of the applied rectified A.C. voltage. After a brief portion of the pulsating DC. voltage transformer 252 saturates and draws sufficient current to charge condenser 250 and maintain conduction through resistor 248. The resultant voltage generated across transformer 252 is a sharp spike of voltage occurring at the appropriate point in each cycle of the pulsating rectified line voltage. This sharp spike of voltage serves to trigger thyratron 256 since the secondary of transformer 252 is phased to cause conduction of thyratron 256. Thus, at a time near the beginning of each of the pulses per second output, thyratron 256 is triggered into conduction. This triggering point occurs When the pulsating voltage output of rectifier 246 exceeds the residual voltage across condenser 250.

During a condition of open circuit, in which the gap between the electrode 202 and workpiece 200 is too great for conduction, a sharp high current surge is drawn through inductance 262 and a corresponding voltage is developed across transformer 264. In this open circuit condition, only the magnetizing current of transformer 264 flows through resistor 270. The time constant of choke 262 and condenser 260 is selected to produce a very brief interval for surge of current flow and a short duration pulse of voltage across transformer 264. As condenser 260 becomes charged to a voltage roughly equal to that stored across condenser network 226, the stored energy in the field of inductance 262 sustains conduction of thyratron 256 overcharging condenser 260. After collapse of the inductive energy in inductance 262, condenser 260 is overcharged to a voltage approximately twice that of condenser 226.

The resultant voltage at that instant around network 226, 256, 262, 260, is such that a negative voltage occurs on the plate of thyratron 256. This negative voltage in accordance with the characteristics of thyratron operation de-ionizes thyratron 256 and thus allows the grid of thyratron 256 to regain control, which at that instant is biased negative by condenser 242. Thus, a sharp pulse of voltage is applied across the working gap from the secondary of transformer 264 through diodes 268; and this sharp high voltage pulse occurs in-phase and corresponds to the power pulse from transformer 204 and the appropriate power network.

Should the gap ionize, and conduction result, current is drawn through diode 268 and secondary of transformer 264. This current flow is limited by resistor 270 series connected in the primary of transformer 264 to a value in accordance with the design. Should the working gap be shorted and result in no voltage across transformer 264, resistor 270 has sufiicient resistance to allow choke 262 to function properly by overcharging condenser 260. Were it not for resistor 270, extremely high damaging currents would flow through thyratron 256 and no current would flow through choke 262 and in this instance, condenser 260 would not overcharge. Since a negative Volt- -age would not be applied to the plate of thyratron 256 during this failure condition, D.C. conduction would result and the grid of thyratron 256 would have no control. Once D.C. conduction resulted, the machine would have to be shut off to allow the grid to regain control. Thus, the trigger pulses for high striking voltage pulses would disappear. For this reason, it is essential to achieve the proper balance between condenser 260, choke 262 and resistor 270. It is only necessary that resistor 258 be of such magnitude to limit the rate of discharge of condenser 260. Typical surge conditions through thyratron 256 would include a peak current of 300 amperes and an average or DC. current of 5 amperes. Through use of storage condenser 226, greatly increased current flow results during the striking voltage pulse for short duration than would otherwise be possible. In this manner, power economy is achieved since the striking voltage pulse lasts for a very brief portion of each cycle and the bulk of the machining power is delivered through transformer 204 at a low voltage level, just high enough to sustain conduction of the desired magnitude through re sistor network 272 and to limit current flow through the working gap on short circuit, since it is readily apparent that no short circuit current control exists. This circuit has one outstanding advantage not possible in any of the other forms of superimposed high striking voltage and that is through use of thyratron 256 and the triggering network, not only very high voltage may be generated but also high current during the brief interval of the striking voltage pulse. The high power pulse helps maintain the gap even at an increased distance since it disintegrates any medium impedance stringers or short circuits that may tend to bridge the working gap and observation of its performance indicates somewhat better reliability than the other high voltage, high impedance triggering circuits. 'This particular circuit is limited to twice the frequency of the input A.C. voltage and also to the relatively low frequency of operation for the thyratron which under no circumstances could exceed 1000 cycles per second for most commercially available xenon filled thyratron It must be noted that the A.C. voltage developed across secondary 212 of transformer 204 is of relatively low magnitude compared to the triggering voltage developed from the thyratron circuit. Full wave rectifier assembly 214 must be of suflicient voltage rating to block or Withstand the inverse voltage developed from the triggering pulse. In other words, this voltage is applied and divided across balancing resistors 216 such that approximately half the voltage of the triggering voltage occurs across each resistor and in this manner the power from the superimposed circuit is blocked from secondary 212 of the power transformer by this high voltage rectifier net work in a manner very similar to the circuitry of FIG- URE 1. The triggering pulse must be phased by the choice of resistor 248, condenser 250 to occur sometime after the rectified sinusoidal voltage output of rectifier 214 is high enough to sustain conduction through the arc.

Attention is now directed to FIG. 10, which is a circuit employing transistors for higher frequency operation and a novel network for achieving the superimposed high striking voltage. The novel network for achieving the superimposed high striking voltage comprises a step-up transformer 358 and associated components about to be described. FIGURE 10 is a circuit embodying transistors for the control of the pulsating are power as well as in the preamplifier. It is essential to realize that in this instance, rectangular pulses are also generated in a manner very similar to the circuitry of FIGURE 1. In the transistor circuitry of FIGURE 10, the working gap consisting of electrode 302 and workpiece 300 is connected through dropping resistor 308, to the collector of transistor 306. The emitter of transistor 306 is connected to the positive terminal of the E.D.M. DC. power supply 304. The pulser amplifier for output transistor bank 306 is similar at least in principle to the circuitry of FIGURE 1. Transistor 306 is generally many transistors, perhaps hundreds of transistors capable of generating the very hi h output machining currents required in E.D.M. PNP transistor 314 may represent a bank of transistors for the pre-amplifier in a manner analogous to that of the tube bank 76 in the circuitry of FIGURE 1. In this circuitry, transistor driver bank 314 is nonconductive during conduction of transistor 306. PNP

type transistor 306 is rendered conductive by DC. power supply 312 through resistor 326 and choke 324. Conduction of transistor driver bank 314 connects the base of power bank 306 and shunts the current flow from resistor 326 and choke 324, such that the direction of electron fiow in this instance is from drive voltage 312 through resistor 326, choke 324, collector-emitter of transistor 314 and back to the positive terminal of voltage 310.

Drive current during ON time of transistor 306 is furnished from battery 312-through resistor 326, choke 324 and the base emitter circuit of transistor 306 back to the positive terminal of voltage 312. Choke 324 as well as choke 318 and 330 are included to provide sharp leading edge drive of the appropriate transistor network. During a period of conduction of transistor 314, increased electron fiow is drawn through resistor 326 and choke 324 in accordance with the higher total voltage of bias 310 and drive voltage 312. As transistor 314 shuts off instantaneously, this increased. electron flow is forced or accelerated through the base emitter circuit of transistor 306, thus providing sharp leading edge drive in an accelerated manner for the duration of the inductive efiect of choke 324. In a similar manner, as transistor 314 becomes instantaneously conductive, the increase in electron fiow through choke 324 is momentarily retarded and provides for a sharp cut-off pulse to transistor 306, thus assuring vertical rise and fall and sharp switching action of each particular transistor stage. Similarly, NPN transistor 320 drives transistor 314 drawing electron flow from drive supply 312 through bias resistor 322, emitter collector of transistor 320 and base emitter circuit of transistor 314. Electron flow is momentarily retarded through choke 318 thus providing a sharp surge to transistor 314 for turn ON through base emitter circuit of transistor 314 and bias resistor 350. During conduction, a shunt electron flow also occurs through choke 318 and resistor 316. As transistor 320 is switched OFF sharply, choke 31S sustains electron fiow in the same direction and sharply cuts off transistor 314 causing cut-off electron tlow through resistor 316, resistor 350 and clearing the emitter base circuit of transistor 314.

NPN transistor 320 is likewise rendered conductive by the first drive transistor shown in this amplifier as transistor 334. Thus electron flow for drive of transistor 320 occurs from the negative terminal of supply 312 through limiting resistor 322, emitter base circuit of transistor 320, collectorcmitter of transistor 334, resistor 332, bias supply 310, to the positive terminal of drive voltage 312. After a short delay determined. by inductance 330, a shunt electron flow is also drawn through resistor 328 and inductance 330 in parallel with network 322, 320. As transistor 334 shuts off sharply, choke 330 sustains a cut-off electron flow through base-emitter of transistor 320, resistor 322, resistor 328, thereby clearing and sharply cutting off transistor 320.

The pulser drive shown in this instance as pulser 336 may be a tube type of pulser or multivibrator as shown in FIGURE 1, or it may be a commercially available pulser of suitable characteristics, or it may be a transistor multivibrator designed for particular control of the circuitry.

In a manner very similar to that of FIGURE 1, transistor 338 operates as a per-pulse cut-cit device in the circuitry of FIGURE 10. It must be noted, in this instance, that when transistor 334 is rendered conductive, output transistor bank 306 is rendered nonconductive. The machining pulse in FIGURE 10 occurs when transistor 306 is conductive and. is interrupted during normal operation by the conduction of pulser 336 at selected time intervals through the base-emitter circuit of transistor 334.

Prior to the start of a machining pulse, pulser 336, transistor 334, transistor 314, are all conductive biasing power transistor bank 306 OFF. In this condition, transistor 338 is aso biased OFF by the absence of any drive signal in its base circuit and by virtue of the direct resistance connection from the base of transistor 338 through potentiometer arm 340 and the lower leg of potentiometer 344 through the lower portion of potentiometer 346 to the emitter of transistor 338. Since no voltage exists in this loop, cut-01f transistor 338 is nonconductive. At the initiation of a machining power pulse, pulser 336 becomes sharply nonconductive, rendering transistor 334 and transistor 314 nonconductive, thus per '1 1 mitting conduction of power transistor 306. If the space between electrode 302 and workpiece 300 is sufficient to permit voltage across the working gap, this voltage is also presented across potentiometer 346 and a portion of this voltage is presented at tap 348. After a momentary delay interval determined by the relative magnitude of condenser 342, the upper portion of potentiometer 344 and resistor 352, a keying signal occurs at potentiometer arm 340. The per-pulse cut-oft" operation, in this instance, compares the relative magnitude between the portion of the arc voltage at 348 and the keying signal at 340. If the arc voltage is of a sufficient magnitude to overcome the voltage at tap 340, transistor 338 is maintained in a nonconducting condition and thus does not affect the operation of the power circuitry. If the voltage at tap 348 is less than that of keying reference 340, transistor 338 becomes instantaneously conductive with drive electron flow in this instance occurring from the negative termi- -nal of power voltage 304 through the lower portion of potentiometer 346, potentiometer arm 348, emitter-base of transistor 338, the upper portion of potentiometer 344 and resistor 352, collector-emitter of transistor 306, balancing resistor 354 to the positive terminal of power voltage 304, thus rendering transistor 338 conductive. This condition of conduction corresponds exactly to the performance of the other circuits in which a voltage lower than the preset magnitude occurring across the arc will instantaneously render the cut-off device active. In this instance, conduction of transistor 338 drives transistor 334 in such a manner as to interrupt conduction of transistor 306, thus instantaneously squelching the faulty pulse in the output.

In a manner similar to that of the previous circuits, transistor 338 may be so connected to directly affect the operation of the pulser by triggering the multivibrator portion of that pulser. In the circuitry shown in FIG- URE 10, however, cut-off transistor 338 overcomes the action of pulser 336 and operates independently of the pulser to shut off the faulty cutting power. Performance of the circuitry in this manner has the one advantage that after the very short delay time encountered in the tran- I sistor components and the various stages of the amplifier, it is possible to re-ignite the are immediately without waiting for the normal interval between pulses caused by pulser 336. Of course, no pulse of duration longer than that determined by pulser 336 is permitted and the action of the cut-off transistor 33% in this instance is only to cut-off the faulty portion of any particular pulse.

The gap between electrode 302 and workpiece 300 is fed through transistor 306 which represents a bank of transistors the number of which will be dictated by the power requirements of the machine.

It may be seen from FIG. that transformer 358 is connected to power voltage 304 through condenser 364,

resistor 308 and power transistor bank 306 and limiting resistor 354. The secondary of transformer 358- is wound and phased such that a voltage boost is achieved from the fundamental machining power voltage 304 of the desired increase generated on the secondary of transformer 358. Diode 360 connected to the output side of the secondary of transformer 358 blocks reverse conduction through transformer 358. In a manner similar to the triggering of transformer 252 shown in FIG. 9, transformer 358 saturates sharply after the initial pulse of voltage occurs for triggering and the resultant output wave form would be identical to that of 189 of FIG. 13. Diode 356 blocks the superimposed high striking voltage from resistor 308 and transistor bank 306. In addition, during the superimposed high striking voltage, rectifier 368 becomes conductive through resistor 366 to clip the superimposed spike of voltage from rheostat 346.

Should the electrode spacing be sufficient to ionize the gap, sufficient current is drawn through transformer 358 to charge condenser 364 such that no additional current fiow occurs beyond the initial leading triggering spike.

The charging time of condenser 364 is short with respect to the total pulse duration so that only a small amount of power is lost through the less efficient high voltage circuitry. The time constant of condenser 364, resistor 362, is of the order of time of the OFF time between pulses, such that condenser 364 discharges substantially between cycles.

From this analysis, it becomes apparent that the common denominator of all superimposed high voltage circuits consists of the provision for the high striking voltage and a blocking diode of a voltage rating sulficient to block this high striking voltage and a current rating sufficient to carry full machining current. In the circuitry of FIGURE 10, this diode is 356; in circuitry of FIGURE 9, the diode is 214; and so on for each specific instance.

An additional form of circuitry embodying superimposed high striking voltage is an RC type of circuit or relaxation oscillator circuit, embodying the same principle of operation. An example of this type of circuit is shown in FIGURE 11. In the superimposed high striking voltage RC circuit shown in FIGURE 11, DC. supply voltage 404, resistor 406 and condenser 408 form the fundamental high current RC circuit for electrode 402 and workpiece 400. Resistor 406 and condenser 408 would be selected in accordance with normal design procedures for RC circuitry.

The superimposed high voltage portion is obtained from high D.C. supply voltage 416, resistor 414 and condenser 410. It may be seen that upon initiation of any charging cycle, condenser 408 is charged through resistor 406 by supply voltage 404 and superimposed condenser 410 is charger through resistor 414 by supply voltage 416. During the charging cycle, diode 412 blocks conduction from supply voltage 416. The total voltage impressed across the working gap is therefore the voltage stored across condenser 408 added to the superimposed voltage across condenser 410. Should the gap distance be short enough, such that a discharge results, condenser 410 is rapidly discharged through the machining gap and the balance of the current from condenser 408 discharges through the machining gap and through diode 412, thus completing one complete discharge cycle of this relaxation oscillator form of operation. As in each case with the superimposed circuitry, tremendous economy occurs, since battery 404 need only be high enough voltage to sustain the arc in accordance with normal principles of design of an RC type of circuit, and battery 416, which is the superimposed high striking voltage, need have only sufficient power to ionize the are at the high voltage level since the bulk of machining power is supplied by DC. power source 404. A circuit of this type is virtually mandatory where the benefits of high striking voltage are to be achieved with an RC power supply, since for any substantial machining power, extreme limit would be placed not only on the input power supply 404 but on the power dissipating capacities of resistor 406. Typical machining currents are in excess of amperes and a superimposed high striking voltage in excess of 300 volts which would result in 30 kilowatts of wasted power in resistor 406. The size and cost of such a resistor suitable for an RC type E.D.M. circuit would be prohibitive but would become entirely practical when this method of superimposed circuitry is employed. It is essential for proper operation of this circuit that the charging time constant of 408- 406 be less than the time constant of 410-414 so that triggering occurs only when condenser 408 is substantially charged.

It will be understood from the above that the provision of a superimposed high striking voltage in connection with the regular gap voltage in E.D.M. apparatus results, among other advantages, in greater stability of power feed operation because it permits the use of a wider gap. By making the gap spacing respond to the higher striking voltage rather than the lower machining voltage, the wider gap is maintained yet this wider gap is readily ionized without high expenditure of power and the finish and accuracy of cut are not impaired.

In the foregoing examples, the means for passing power pulses across the gap is shown as either a bank of vacuum tubes or a bank of transistors. This electronic switch may be any device having a circuit-controlling or gating ability by virtue of which a pulsating signal of relatively small magnitude is capable of triggering the switch at high frequency without the use of mechanical devices such as relays.

I claim:

1. In an electrical discharge machining apparatus for eroding a conductive workpiece by intermittent electrical discharge across a gap between an electrode and a workpiece in the presence of a dielectric coolant, a source of machining voltage comprising an impedance matching transformer having its primary connected to a power source and its secondary connected across said gap, a condenser connected in series with said secondary and said gap, a one-way current conducting device connected across said condenser, and a one-way current conducting device connected across said gap, whereby the flyback voltage between gap discharges is stored across the condenser and is effective to ionize the gap to initiate the following discharge.

2. In an electrical discharge machining apparatus for eroding a conductive workpiece by intermittent electrical discharge across a gap between an electrode and a workpiece in the presence of a dielectric coolant, a source of machining voltage comprising a transformer having its primary connected to an AC. power supply and its secondary connected across the gap, a second source of voltage of higher magnitude than said first source, said second source comprising a transformer having its secondary connected across said gap and its primary connected to a DC. power supply through a thyratron biased normally non-conductive, and a trigger network associated with said thyratron for firing said thyratron in timed relation with said first voltage source.

3. In an electrical-discharge machining apparatus for eroding a conductive workpiece by intermittent electrical discharge across a gap between an electrode and a workpiece in the presence of a dielectric coolant, a power supply, means for pulsing said power supply thereby to produce a series of timed, short duration, power pulses, and means coupling said power supply to the gap comprising a matching transformer having its primary winding connected to said power supply, said transformer having a first secondary winding connected across said gap in series with a rectifier and a resistor and a second secondary winding of lower voltage output connected across said gap through a rectifier.

4. In an electrical discharge machining apparatus for eroding a conductive workpiece by intermittent electrical discharge across a gap between an electrode and a workpiece in the presence of a dielectric coolant, a source of DC. power, an electronic switch capable of being rendered alternately conducting and non-conducting at relatively high frequency, a pulser for pulsing said switch, a transformer, circuit means connecting the primary winding of said transformer to said power source through said switch, said transformer having a first secondary winding connected across said gap through a one-way current conducting device and a resistor in series, and a second secondary winding of difierent voltage output connected across said gap through a one-way current conducting device.

5. The combination of claim 4 wherein the voltage output of said first secondary winding is of higher magnitude than that of said second secondary winding.

6. The combination of claim 4 wherein the one-way current conducting device connecting the lower voltage Winding to the gap is of sufiicient current rating to carry the full rated machining current of said lower voltage winding and of sufficient voltage rating to block power flow from the higher voltage winding.

7. In an electrical discharge machining apparatus for eroding a conductive workpiece by intermittent electrical discharge across a gap between an electrode and the workpiece in the presence of a dielectric coolant, a source of DC power, a transformer, means connecting the primaryof said transformer to said power source, pulsing means connected between said power source and said primary for pulsing said primary at selected frequency, said transformer having two secondary windings of different voltage output, means connecting the secondary of lower voltage output across said gap through a one-way current conducting device, and means connecting the secondary winding of higher voltage output across said gap through a network comprising a one-way current conducting device, a series connected resistor and a condenser shunted across said resistor, whereby upon pulsing of said primary, a low-impedance striking voltage pulse is delivered from said high voltage network which rapidly ionizes the gap and permits a relatively high power pulse to be delivered from said low voltage winding.

8. The combination set forth in claim 2 in which said trigger network is comprised of a saturable transformer having its primary connected in series with an RC network consisting of a capacitor and shunt resistor and a rectified A.C. voltage derived from said AC. power supply.

9. The combination set forth in claim 8 in which said saturable transformer is designed to saturate in a relatively small period of the rectifier voltage exceeding the residual voltage stored across said capacitor.

10. The combination set forth in claim 9 in which the RC time constant of said resistor and capacitor is such that rectified voltage exceeds the voltage of said capacitor each cycle after the voltage of said first mentioned secondary exceeds the magnitude required to sustain gap conduction.

11. In an electrical discharge maching apparatus for eroding a conductive workpiece by intermittent electrical discharge across a gap between an electrode and the workpiece in the presence of a dielectric coolant, a source of maching voltage comprising a source of power and a pulse transformer having its primary connected to said source and its secondary connected across said gap, said pulse transformer having high power, low impedance in one polarity and low power, high impedance in the flyback polarity, a periodically operated electronic switch connected intermediate said source and said primary, a condenser connected in series with said secondary and the gap, a one-way current conducting device connected across said condenser and phased to conduct power in phase with said switch, and a one-way current conducting device connected across the gap, whereby the pulse transformer fiyback voltage between gap discharges is stored across the condenser and effective to ionize the gap to initiate the following discharge.

12. The combination as set forth in claim 11 in which said condenser is of a relatively small capacitance to prevent excessive loading of fiyback voltage.

13. The combination as set forth in claim 3 in which said resistor is of relatively high resistance value and a condenser is connected thereacross to provide increased current flow during the initial portion of the pulse across the gap.

14. A pulse generator for a discharge device of nonlinear load characteristics comprising a first source of voltage including a transformer having its primary connected to an alternating current power supply and its secondary connected across the device, a second source of voltage of higher magnitude than said first source, said second source comprising a transformer having its secondary connected across the device and its primary connected to a direct current power supply through a switching means biased normally nonconductive, and a trigger netwprk operatively connected to and controlling said switching means for firing it in timed relationship with said first voltage source.

15. The combination as set forth in claim 14 in which said switching means comprising a thyratron and said trigger network cornprises a saturable transformer having its primary connected in series with a parallel resistor-condenser network, and a direct current power supply derived from said alternating current power supply.

16. The combination as set forth in claim 15 in which said saturable transformer is operable to saturate in a relatively smaller period of the rectified voltage exceeding the residual voltage stored across said condenser.

17. The combination as set forth in claim 14 in which said first voltage source and said trigger network are operated by a common alternating current supply.

18. A pulse generator for a discharge device having nonlinear load characteristics comprising a source of direct current potential, a transformer having its primary connected to said source, pulsing means connected between said source and said primary for pulsing said primary at selected frequency, said transformer having two secondary windings of different voltage output, means connecting said secondary said secondary of lower voltage output across the device through a one-way carrying element, and means connecting said secondary of higher voltage across the device through a network comprising a one-way current conducting element, a series connected resistor and a condenser shunted across said resistor.

19. In an electrical discharge machining apparatus for machining a conductive workpiece by intermittent electrical discharge across the gap between an electrode and the workpiece in the presence of a dielectric coolant, a source of alternating current potential connected across said gap, a condenser connected in series with source and said gap, a one-way current conducting device conneted across said condenser and phased for conduction in one direction across said gap, and a one-way current conducting device connected across said gap and phased for conduction in the opposite direction across said gap.

20. A pulse generator for a discharge device of nonlinear load characteristics comprising a source of alternating current potential connected across said device, a condenser connected in series between said source and said device, a rectifier connected across said condenser and phased for conduction in one direction across said device, and a rectifier connected across said device and phased for conduction in the opposite direction across said device.

21. A pulse generator for a discharge device of nonlinear load characteristics comprising a source of power and a pulse transformer having its primary connected to said source and its secondary connected across the device, said pulse transformer having high power, low impedance in one polarity and low power, high impedance in the flyback polarity, a periodically operated electronic switch connected intermediate said source and said primary, a condenser connected in series with a said seccondary and the device, a. rectifier connected across said condenser and phased to conduct power in phase with said switch, and a rectifier connected across said device having a phasing to conduct power across said device out of phase with said switch.

22. The combination as set forth in claim 21 in which said condenser is of a relatively small capacitance to prevent excessive loading of flyback voltage.

References Cited by the Examiner GEORGE N. WESTBY, Primary Eraminer. RALPH G. NILSON, Examiner.

UNITED STATES PATENT OFFICE CERTIFICATE OF CORRECTION Patent No. 3,229,159 January 11, 1965 Robert S. Webb It is hereby certified that error appears in the above numbered patent requiring correction and that the said Letters Patent should read as corrected below.

Column 1, line 36, for "suply" read supply column 5, line 12, after "voltage" insert developed line 40, after "tubes" insert 22 column 7, line 35, for "of" read or column 12, line 37, before "across" insert stored column 14, line 12, after "secondary" insert winding line 31, for rectifier" read rectified column 15, line 25, strike out "said secondary", second occurrence; line 26, after "one-way" insert current Signed and sealed this 13th day of December 1966.

SEAL) Ittest:

lRNEST W. SWIDER EDWARD J. BRENNER Meeting Officer Commissioner of Patents 

1. IN AN ELECTRICAL DISCHARGE MACHINING APPARATUS FOR ERODING A CONDUCTIVE WORKPIECE BY INTERMITTENT ELECTRICAL DISCHARGE ACROSS A GAP BETWEEN AN ELECTRODE AND A WORKPIECE IN THE PRESENCE OF A DIELECTRIC COOLANT, A SOURCE OF MACHINING VOLTAGE COMPRISING AN IMPEDANCE MATCHING TRANSFORMER HAVING ITS PRIMARY CONNECTED TO A POWER SOURCE AND ITS SECONDARY CONNECTED ACROSS SAID GAP, A CONDENSER CONNECTING IN SERIES WITH SAID SECONDARY AND SAID GAP, A ONE-WAY CURRENT CONDUCTING DEVICE CONNECTED ACROSS SAID CONDENSER, AND A NON-WAY CURRENT CONDUCTING DEVICE CONNECTED ACROSS SAID GAP, WHEREBY THE FLYBACK VOLTAGE BETWEEN GAP DISCHARGES IS STORED ACROSS THE CONDENSER AND IS EFFECTIVE TO IONIZE THE GAP TO INITIATE THE FOLLOWING DISCHARGE. 